TELKOM
NIKA Indonesia
n
Journal of
Electrical En
gineering
Vol. 12, No. 10, Octobe
r 20
14, pp. 7160
~ 716
6
DOI: 10.115
9
1
/telkomni
ka.
v
12i8.634
5
7160
Re
cei
v
ed Ma
y 30, 201
4; Revi
sed
Jul
y
1
1
, 2014; Acce
pted Jul
y
29,
2014
Higher Efficiency Switching Mode Power Amplifier
Design using the Third-Harmonic Peaking Turning
Mode
NSHU. Victor*, Z. WenBi
a
o, M.R. Anjum
Schoo
l of elect
r
onics a
nd Info
rmation,
Bei
jin
g Institute of
T
e
chn
o
lo
g
y
,
Z
hong
gu
ancu
n
Nand
aji
e
no.5,
Beijn
g,10
008
1
Chin
a
*Corres
p
o
ndi
n
g
author e-m
a
il
: nshuvico
01@
gmail.c
o
m
A
b
st
r
a
ct
T
h
is p
a
p
e
r pr
ovid
es th
e
de
sign
ap
pro
a
ch
of
one
stag
e
sw
itching-
mo
de
bas
ed
clas
s F
pow
e
r
amplifi
e
r (PA).
T
he d
e
vic
e
’
s
non
lin
ear
be
ha
vior w
a
s
an
aly
z
e
d
to
re
duc
e
the
diss
ipate
d
p
o
w
e
r ov
er
th
e
active
devic
e
and
therefor
e
the PA
efficie
n
cy w
a
s incr
e
a
se w
i
tho
u
t h
a
vin
g
to c
o
mp
romise th
e p
o
w
er
amplifi
e
r
’
s si
z
e
. T
he load har
mo
nics w
e
re control
l
ed so
th
at the drain vo
ltage a
nd the
drai
n current
d
o
rarely co
inci
de
w
i
th each oth
e
r, thus gre
a
tl
y incr
eas
e the
pow
er perfor
m
a
n
ce
of the
devic
e. T
a
king
th
e
devic
e si
z
e
a
n
d
cost para
m
eters into co
nsid
eratio
n,
the de
sign of l
oad
ha
rmo
n
ic trap cir
c
uit w
a
s reduc
e
d
only
to th
e 2
nd
and
3
rd
har
mon
i
cs. T
he Ga
N
HEMT
transist
o
r, for its
hi
gh-
spee
d sw
itchi
n
g a
b
il
ity, is
use
d
i
n
our desi
gn to max
i
mi
z
e
the
output pow
er a
nd the opti
m
i
z
ed circuit op
er
ati
ng at 5.8GH
z
w
i
th o
u
tput p
o
w
e
r
of 50.98W
w
a
s simu
late
d. A pow
er adde
d efficiency
of 60%
w
i
th the pow
er gai
n of 14dB w
a
s obtai
ne
d.
Ke
y
w
ords
:
cla
ss-F
,
loadPul
l, har
mo
nic trap, RF
pow
er ampl
ifier, third-h
a
r
m
onic p
eaki
n
g
Copy
right
©
2014 In
stitu
t
e o
f
Ad
van
ced
En
g
i
n
eerin
g and
Scien
ce. All
rig
h
t
s reser
ve
d
.
1. Introduc
tion
The radio f
r
e
quen
cy (RF
)
power
ampli
f
ier ar
e on
e
of the mo
st importa
nt ele
m
ent in
transmitter u
n
its
of co
mm
unication systems.
Th
e
re
quire
ment
s o
n
the
po
wer amplifie
r h
a
v
e
become mo
re
important re
gard
ed to the
system
pe
rfo
r
man
c
e. Fo
r
good p
e
rfo
r
m
ance, not onl
y
a
con
s
id
era
b
le
gain
with hi
g
h
efficien
cy i
s
req
u
ire
d
b
u
t
also
a
suitabl
e output
po
wer i
s
exp
e
cte
d
to
be provid
ed. For the reliabl
e transmissio
n the PA output powe
r
mu
st be sufficient
.
As co
mmuni
cation ma
rket gro
w
ing
up,
digita
l sig
nal
pro
c
e
ssi
ng m
odule
s
an
d the ra
dio
freque
ncy int
egrate
d
ci
rcui
ts (RFIC)
are
usually
inte
g
r
ated into p
o
rtable ele
c
tro
n
i
cs in
ord
e
r t
o
ensure
multi
m
edia
appli
c
ations in
sm
a
ll si
zed
dev
i
c
es. Li
nea
r
RF po
we
r a
m
p
lifiers be
cau
s
e of
their lo
w p
o
w
er efficie
n
cy they have
som
e
d
r
a
w
backs a
s
the
y
con
s
um
e l
a
rge
amo
unt
s of
energy, dissi
pate great h
eat, and o
c
cupy big sp
ace in base st
ations. Cell-p
hone
s an
d o
t
her
portabl
e com
m
unication d
e
vice or
ba
se
-station
s
are
i
n
an
in
creasi
ng p
u
rsuit
of
the efficie
n
cy
to
satisfy the re
quire
ment of long sta
ndby time
and lo
w co
st, all these requi
rem
e
n
t
s depe
nd on
the
power am
plifier in the
s
e d
e
vice
s. So the high effi
cie
n
cy PA dra
w
s mu
ch mo
re
attentions in
this
research field. Signific
a
ntly more effic
i
ent PA te
ch
nol
ogy is ne
ce
ssary to the
evo
l
ution of m
obi
le
system
s. To
achi
eve hig
h
efficien
cy i
t
is ne
ce
ssa
r
y to minimi
ze lo
sse
s
which
are l
a
rg
ely
dissipate
d
in the active de
vice esp
e
ci
all
y
when
the a
m
plifier is op
erating at hig
h
current an
d
voltage level
s
. The
po
we
r dissipate
d
in
the a
c
tive d
e
vice in
crea
ses a
s
th
e ov
erlap
of volta
g
e
and cu
rre
nt
waveforms
i
n
cre
a
ses.
T
o
deal with
t
he power dissip
ated
thou
ght the
active de
vice
we ne
ed to e
n
su
re that th
ere i
s
non
-ov
e
rlap
pi
ng
regi
on of the cu
rrent and volta
ge wavefo
rm
s at
the a
c
tive de
vice; on
ce
thi
s
con
d
ition i
s
rea
c
he
d the
highe
r p
o
wer
efficien
cy can
be
obtain
ed.
In
this pa
per we
use
d
the n
o
n
-line
a
r
device. PAs
ope
ra
ting in the
switch-m
ode
do
main exploit t
he
nonlin
ear
reg
i
on of the de
vice to impo
se a hi
ghly eff
i
cient set of non-
overl
appi
ng cu
rrent a
nd
voltage d
r
ain
waveforms [1
]. The drain v
o
ltage
wavef
o
rm
will b
e
shape
d to a
square
wave a
n
d
curre
n
t wavef
o
rm
will be
shape
d to a h
a
lf-sin
usoi
dal
wave. The
r
e
f
ore, acro
ss the a
c
tive device
there
will be
no overl
ap b
e
twee
n drain
voltage and
cu
rre
n
t wavef
o
rm
s, as
a re
sult, ze
ro p
o
w
er
dissipatio
n i
s
cre
a
ted
whil
e
achieving
in
the same
time the th
eo
reti
cal
100%
dra
i
n efficie
n
cy.
In
pra
c
tice, to
simplify the circuit de
sign
o
n
ly the 2nd a
nd 3rd ha
rm
onics have
b
een con
s
ide
r
e
d
sin
c
e the nu
mber of ha
rm
onics that ca
n be effective
l
y controlle
d is finite.
Evaluation Warning : The document was created with Spire.PDF for Python.
TELKOM
NIKA
ISSN:
2302-4
046
High
er Efficie
n
cy Switchi
n
g
Mode Power
Am
plifier Design u
s
in
g… (NSHU. Victor)
7161
The main p
u
rpo
s
e of th
is pap
er is
to provide a
desig
n pro
c
ed
ure fo
r simplified
broa
dba
nd hi
gher frequ
en
cy Cla
s
s-F p
o
we
r amplifie
r de
sign. In th
is de
sign the
Advance
syst
em
desi
gn (A
DS) softwa
r
e
wa
s u
s
ed
alon
g
the de
sign
p
r
ocess and si
mulation
s
fro
m
the
tra
n
si
stor
modelin
g to the final circuit
optimization.
2.
Class
-F PA
Architec
ture
and Design
Techniqu
es
2.1.
S
w
i
t
ching M
ode Po
w
e
r Amplifier
Architecture
To
sub
s
tantia
lly increa
se
th
e efficie
n
cy
of
a
cla
s
s F
po
wer am
plifier,
we
n
eed
to
p
r
odu
ce
an open
circu
i
t at the odd harmo
nics and
the short ci
rcuit at even harmoni
cs
at the active device
output. With
su
ch requi
red
harmo
nic tu
n
i
ng, the
simul
t
aneou
s ap
pe
aran
ce
of the voltage acro
ss
the a
c
tive de
vice o
u
tput a
nd
curre
n
t through
it
contai
ning th
e h
a
rmonics
of the
sam
e
o
r
d
e
r i
s
to
be avoided.
By analyzing
Fourie
r se
ri
es expan
si
o
n
of the volta
ge and curre
n
t waveform
s as
expre
s
sed i
n
the Equ
a
tion (1
) a
nd
(2) re
spe
c
ti
vely, we
ca
n defi
ne the
output
imped
an
ce,
the
con
d
ition at whi
c
h the po
wer di
ssipate
d
thr
oug
h the active device is minimi
zed and then
the
maximum pra
c
tical d
r
ain ef
ficien
cy can b
e
rea
c
he
d.
1
3,
5
,
7
,
.
.
.
si
n
s
i
n
DD
n
n
VV
V
V
n
(1)
1
2,
4,
6
,
.
.
.
si
n
s
i
n
on
n
I
II
I
n
(2)
Mathemati
c
al
ly taking
an
i
n
finite nu
mbe
r
of
ha
rmoni
cs into
a
c
coun
t, the obtai
ne
d shap
e
of waveform
s is a squ
a
re wave (o
dd h
a
rmo
n
ic) and
a half sinu
soid wave (even harmoni
c) for
the voltage or current resp
ectively.
The
waveforms
sho
w
n i
n
Figure 1
re
su
lt in 100% of
drain
efficien
cy. Whe
n
we
come
to
the pra
c
tice
in the re
al
worl
d, to co
ntrol imp
eda
nce
s
for
an
infinite numb
e
r of ha
rmo
n
ics
become
s
very difficult.
Figure 1.
Wa
veforms
whi
c
h Re
sult in an
Efficiency of 100%
Figure
2
.
The
Drain
Wavef
o
rm
s in Pra
c
tice
Evaluation Warning : The document was created with Spire.PDF for Python.
ISSN: 23
02-4
046
TELKOM
NI
KA
Vol. 12, No. 10, Octobe
r 2014: 716
0
– 7166
7162
Greate
r
the
numbe
r of h
a
rmo
n
ics i
s
, better t
he effi
cien
cy is. To
achi
eve a n
o
ticea
b
le
increa
se i
n
ef
ficien
cy we n
eed
a hi
ghe
r
numbe
r
of
ha
rmoni
cs al
so
to be ta
ke
n i
n
to a
c
count
as
the Figu
re 2
shows [2], but
as
we in
crea
se the
nu
mbe
r
of harm
oni
cs the ci
rcuit be
come
comple
x
and th
e final
device
si
ze
i
s
al
so
en
gag
ed, for th
is reason
we
typically
u
s
e
the
Third-Ha
rmo
n
ic
pea
king
[3, 7]
tuning
mod
e
in which the
i
nput imp
eda
n
c
e
of the
outp
u
t network i
s
only controll
e
d
up to third ha
rmoni
c.
The o
u
tput n
e
twork
de
sig
n
sh
ould
also involv
e the
accepta
b
le
matchin
g
circuit at the
fundame
n
tal
freque
ncy to
achi
eve hig
h
effici
en
cy. The two fu
n
damental
bal
ance conditio
n
s
sho
u
ld be satisfied to achi
eve 100% drain efficien
cy (
d
): as de
scribed in the E
quation (3), the
power di
ssip
ations
at the
highe
r ha
rmo
n
ic fr
eque
nci
e
s m
u
st b
e
zero
and th
e
DC
po
wer su
pply
must be eq
ua
l to the power generated at
the fundame
n
tal freque
ncy.
1
100%
d
DC
P
P
(3)
Whe
r
e
P
1
is p
o
we
r gen
erated at the fund
amental fre
q
u
ency.
P
DC
is
D
C
power
supply.
The first
con
d
ition to b
e
realized the
lo
ad im
ped
an
ce at
odd
ha
rmoni
cs i
s
t
herefo
r
e
sup
p
o
s
ed
to
be tun
ed
to
open
circuit
(Z
load
≈
∞
) an
d
the l
oad
im
peda
nce
at
even h
a
rm
oni
c
sho
u
ld be tu
ned short
circuited (Z
load
≈
0). Fo
r the
second
con
d
ition som
e
adju
s
tments
are
carrie
d out for maximizin
g
drain effici
en
cy.
In this metho
d
, the phase
betwe
en voltage and
cu
rrent at the harmonics is al
ways ±90
degree
s, so that the power
factors at all harm
oni
c freq
uen
cie
s
be
co
me ze
ro [4].
2.2.
Class F Po
w
e
r Amplifier
Design Pro
c
edure
CG
H40
045 G
a
N HEM
T
is t
he tran
si
stor
use
d
in ou
r d
e
sig
n
. It is biase
d
as V
D
S
= 28 V,
IDQ =
74
1 mA
and
VG
S=-2.1V (Fig
ure 3),
a sta
b
le
facto
r
of 1.975 wa
s computed
ove
r
ou
r
desi
gn centra
l frequen
cy of 5.8 GHz
(Fig
ure 4
)
.
The
software
used is Advanced Desi
gn
System
. The
following subsections illustrate the
desi
gn proce
dure
we u
s
ed
.
Figure 3. Bias Point Dete
rmination
Figure 4. Stability Factor P
l
ot
as Function of Frequency
5
1
01
52
02
5
0
30
0
1
2
3
4
5
6
7
-1
8
VD
S
I
_
Pr
obe1.
i
,
A
m1
m1
VDS
=
I
_
P
r
ob
e1.
i
=
0
.
741
V
G
S
=
-
2
.
100
00
0
28
.
0
00
5
1
01
52
0
2
53
03
54
04
55
0
0
55
1.
0
1.
5
2.
0
2.
5
3.
0
0.
5
3.
5
in
d
e
p
(
d
B(
S_
St
ab
C
i
r
c
l
e
1
)
)
dB
(
S
_S
t
a
bC
i
r
c
l
e1)
m3
m3
i
n
dep(
m
3)
=
dB
(
S
_S
t
a
bCi
r
c
l
e
1)
=
1
.
106
f
r
eq
=
5
.
8
00000G
H
z
25
i
n
d
ep(
S
_
S
t
a
b
C
i
r
c
l
e1)
(
0
.
0
00
t
o
51
.
0
0
0
)
S_
St
ab
C
i
r
c
l
e
1
m1
m1
i
n
de
p(
m1
)
=
S
_
S
t
a
b
C
i
r
c
l
e
1
=
1
.
45
2 /
6.
15
4
fr
eq
=
5
.
8
00
00
0G
H
z
i
m
pe
da
n
c
e =
Z
0
*
(
-
5.
0
1
2
+
j
1
.
4
0
7
)
5
Evaluation Warning : The document was created with Spire.PDF for Python.
TELKOM
NIKA
ISSN:
2302-4
046
High
er Efficie
n
cy Switchi
n
g
Mode Power
Am
plifier Design u
s
in
g… (NSHU. Victor)
7163
2.3. Harmonic
Net
w
o
r
ks
Arc
h
itec
ture
The commo
n
pra
c
tice in
Cl
ass F po
we
r amplifier d
e
si
gn involves t
he de
sign
of a turnin
g
netwo
rk fo
r lo
ad harmoni
cs basi
c
ally de
signed u
p
to
a certain
ord
e
r harmo
nics a
s
sh
own by the
Figure 5, the desi
gn of mat
c
hin
g
network at t
he fund
amental fre
q
uen
cy is also
engag
ed (Fi
gure
6(b
)). The turning net
work
at the odd order ha
rm
o
n
ic
looks like an
open
circuit, at the even order
harm
oni
cs it
pre
s
ent
s a short ci
rcuit. Takin
g
the
co
mplexity of th
e circuit an
d the possibly l
o
ss
introdu
ce
d by the higher order ha
rmo
n
ics into the
design in con
s
id
eration, in this pape
r we o
n
ly
con
s
id
ere
d
the se
con
d
and
third ord
e
r h
a
rmo
n
ics (Fi
g
ure 6
(
c)).
Figure 5. The
Load Harm
o
n
ic Tuni
ng Network Desi
g
n
Saturated
drain
curre
n
ts,
bre
a
kdo
w
n
voltages
are
some
of the
cha
r
a
c
teri
stics of th
e
radio frequ
en
cy power tra
n
s
isto
rs.
To find the load impe
dan
ce co
rrespon
ding
to the maximum po
wer the two
extreme
values of d
r
ai
n voltage
and
cu
rrent a
r
e
e
ngag
ed, re
sul
t
ing to thei
r e
x
cursion
s
f
r
o
m
nea
r
ze
ro t
o
nearly the ma
ximum values. Using the Smith cha
r
t,
fo
r a given deli
v
ery amount of RF power
we
can find th
at for a sp
ecifi
ed maximum
drai
n voltag
e the co
rrespondi
ng loa
d
impeda
nces lie
along p
a
rall
el
-re
si
stan
ce li
nes. Similarly
for a sp
e
c
ified maximum
current we can find that the
impeda
nces f
o
llow a
seri
es-re
si
stan
ce li
ne.
(b)
(c
)
Figure 6. (b)
First ha
rmo
n
i
c
trap, (c)
Harmonic n
e
two
r
ks
re
spon
se
Con
s
tant-po
w
er
conto
u
r
for an ide
a
l power am
plifier ha
s a foo
t
ball sha
pe. In a real
power amplifi
e
r, d
ue to
the
pa
ra
sitic
ele
m
ents su
ch a
s
the
d
r
ain
capa
citan
c
e
a
nd b
ond
-wi
r
e
or
packa
ge in
du
ctan
ce
we h
a
v
e an em
bed
ded
“virtual
d
r
ain
”
. The
dra
i
n imped
an
ce
tran
sform
a
tion
affects the
consta
nt-po
w
e
r
co
ntou
rs
which b
e
co
me
rotated eve
n
disto
r
ted.
These ad
dition of
se
con
d
-o
rd
er effects, lead
the power
-co
n
tours to the elliptical
shap
e.
Whe
n
de
sig
n
i
ng the
output
matchi
ng
circuit of tr
aditio
nal RF am
plifier, such a
s
L
N
A, we
need to d
e
si
gn an o
u
tput
matchin
g
ci
rcuit which is
t
he conjug
ate
match
with the output of t
he
transi
s
to
r. Becau
s
e conju
g
a
te matche
d circuit le
t the amplifier get
s the maximum gain, but gain
is n
o
t the o
n
ly con
s
id
eration when
we
are
de
si
gn po
we
r a
m
plifier; the
more imp
o
rtant
con
s
id
eratio
n
of power
am
plifier is the
Powe
r A
dde
d
efficien
cy (P
AE) whi
c
h th
e expression
is
given in Equa
tion (4).
O
dd har
m
o
n
i
c
s
bl
oc
k
f
i
l
t
e
r
s
O
pen f
o
r
f
u
n
dam
e
n
t
a
l
s
h
o
r
t f
o
r
ha
r
m
on
i
c
s
DC_
F2
VD
D
DC_
B
2
3r
d
5th
nth
T
C
r
ee
C
G
H
400
45
F
Rl
o
a
d
1s
t
2
345
678
9
1
10
-6
0
-5
0
-4
0
-3
0
-2
0
-1
0
-7
0
0
fr
e
q
,
G
H
z
d
B
(S
(1
,
1
))
m1
m3
d
B
(S
(2
,
1
))
m2
m1
freq=
dB
(S(1,1))=
-
2
.
926
5.380GH
z
m2
freq=
dB
(S(2,1))=
-
1
.
722E
-5
5.800GH
z
m3
freq=
dB(S
(1,1
)
)
=
-
2.826
6.19
0GH
z
6
8
10
12
14
16
18
4
20
-8
0
-6
0
-4
0
-2
0
-1
0
0
0
fr
e
q
, G
H
z
d
B
(S
(1
,
1
))
d
B
(S
(2
,
1
))
m1
m1
fre
q
=
dB(S(
2
,1))
=
-
98
.635
17
.40
G
H
z
Evaluation Warning : The document was created with Spire.PDF for Python.
ISSN: 23
02-4
046
TELKOM
NI
KA
Vol. 12, No. 10, Octobe
r 2014: 716
0
– 7166
7164
ou
t
i
n
DC
P
P
PAE
P
(4)
Whe
r
e Pout
, Pin
and
P
DC
stand, re
spe
c
tively, for RF powe
r
output, input drive po
wer a
nd DC
power supply
.
The tran
sisto
r
p
a
ra
sitic el
ements a
r
e t
he m
a
jor fact
or th
at di
sturbs th
e o
pen
-l
oad
and
the sho
r
t-loa
d
conditio
n
s f
o
r the harm
o
nics, to mi
nimize their eff
e
ct on the de
vice perfo
rma
n
ce
we
used th
e
load
pull te
ch
nique
to find
out the
be
st l
oad im
ped
an
ce
of ou
r
po
wer am
plifier to
operate in th
e maximum
power effici
e
n
cy with a
certain hi
gh g
a
in. Load p
u
l
l
is a tech
niq
u
e
whe
r
ein th
e load imp
edan
ce seen
by the device
und
er test (DUT)
is varie
d
and
the perfo
rma
n
ce
of the DUT i
s
simultan
eou
sl
y measu
r
ed [
5
, 6].
Figure 7
.
The
Best Load I
m
peda
nce of the Amplifier
As the
Figu
re 7
sho
w
s,
we fin
d
that t
o
a
c
hieve th
e de
sired
out
put po
we
r of
50
W, we
found that th
e load imp
e
d
ance co
rresp
ondin
g
t
he re
quire
d outp
u
t powe
r
i
s
20.
877-j
25.38
4; in
this case the
PAE above 60% was obtai
ned.
2.4. Input/output
Net
w
o
r
k
s
M
a
tching a
nd
Design O
p
ti
mization
For the
freq
uen
cie
s
bey
ond 5
00M
Hz desi
gnin
g
the filter
with
discrete co
mpone
nts
become
s
difficult to
re
alize
as the
wavel
ength
and
th
e phy
sical di
mensi
o
n
of t
he filter be
co
me
comp
arable,
this re
sult
s in
variou
s lo
sses
wh
i
c
h ma
y cau
s
e
seve
ral de
gradati
on of the
circuit
perfo
rman
ce.
Thu
s
, in o
r
d
e
r to
reali
z
e
the practi
cal
filter and
net
work m
a
tchi
n
g
ci
rcuit, in this
desi
gn
th
e
lu
mped co
mpo
nent were co
nverted
i
n
to
distrib
u
tion el
ement. We
a
dopt
the
u
s
e
of
micro-strip lin
es.
(a) O
u
tput ci
rcuit optimization
(b) T
he outpu
t impedan
ce
matchin
g
Figure 8. S21 and S11 Parameters
m1
i
n
d
ep(m
1
)=
PAE_
c
o
n
t
o
u
r
s
_
p
=
0
.
477 /
-1
10.
0
9
9
l
e
vel
=
61.10153
3,
num
be
r=
1
i
m
ped
anc
e =
Z
0
*
(0.497
-
j
0
.576)
6
m2
i
n
dep(
m
2
)
=
P
del
_c
ont
our
s
_p=
0.
47
5 /
-
109
.
2
9
5
l
e
v
e
l
=
47.
46478
6,
num
ber
=
1
i
m
ped
anc
e =
Z
0
*
(
0
.
504 -
j
0
.
5
82)
25
i
n
d
ep(
P
A
E
_
c
o
n
t
o
u
r
s
_p)
(
0
.
0
00 t
o
8
2
.
000
)
P
A
E
_
c
o
nt
ou
r
s
_
p
m1
i
n
d
ep(
P
d
e
l
_
c
on
t
o
ur
s
_p)
(
0
.
0
0
0
t
o
9
8
.
0
00)
P
d
el
_
c
on
t
o
ur
s
_
p
m2
m1
i
n
d
ep(m
1
)=
PAE_
c
o
n
t
o
u
r
s
_
p
=
0
.
477 /
-1
10.
0
9
9
l
e
vel
=
61.10153
3,
num
be
r=
1
i
m
ped
anc
e =
Z
0
*
(0.497
-
j
0
.576)
6
m2
i
n
dep(
m
2
)
=
P
del
_c
ont
our
s
_p=
0.
47
5 /
-
109
.
2
9
5
l
e
v
e
l
=
47.
46478
6,
num
ber
=
1
i
m
ped
anc
e =
Z
0
*
(
0
.
504 -
j
0
.
5
82)
25
50.
000
Sy
s
t
e
m
R
e
f
e
r
e
nc
e
I
m
pedanc
e
P
A
E
(
t
hi
c
k
)
and D
e
l
i
ver
e
d
P
o
w
e
r (t
h
i
n
)
C
o
n
t
o
u
r
s
47.
97
Ma
x
i
m
u
m
Po
w
e
r
D
e
l
i
v
e
r
ed,
dB
m
61.
20
M
a
x
i
mu
m
P
o
w
e
r
-
A
dded
E
f
fi
c
i
e
n
c
y
, %
4
.
5
5
.
0
5.
5
6
.
0
6.
5
4.
0
7.
0
-5
0
-4
0
-3
0
-2
0
-1
0
-6
0
0
f
r
eq,
G
H
z
d
B
(S
(2
,1
))
m2
d
B
(S
(1
,1
))
m2
f
r
eq=
dB(
S
(
2
,
1
)
)
=
-
4.
369E-
4
5.
800G
H
z
f
r
eq (
4
.
000G
H
z
t
o
7.
000G
H
z
)
S(
2
,
2)
m1
S(
1
,
1)
m1
f
r
eq=
S
(
2,
2)
=
-
1.
946E
-
5
+
j
0
.
010
i
m
peda
n
c
e =
Z0 *
(
1
.
000 +
j
0
.
0
20)
5
.
800G
H
z
Evaluation Warning : The document was created with Spire.PDF for Python.
TELKOM
NIKA
ISSN:
2302-4
046
High
er Efficie
n
cy Switchi
n
g
Mode Power
Am
plifier Design u
s
in
g… (NSHU. Victor)
7165
After designi
ng the harm
o
nic networks we nee
d to match the ne
twork de
sign
ed to the
load impe
da
nce
com
pute
d
by the load
pull simul
a
tion. This i
s
re
alize
d
in two
steps. Th
e first
step i
s
to fin
d
the im
ped
a
n
ce
form
ed
by the loa
d
of 50
Ω
a
nd t
he h
a
rmo
n
ic
netwo
rks
at the
output n
e
two
r
k. The
value
of imped
an
ce
obtaine
d i
s
4
6
.629
+j0.30
5
2
.
In the
seco
nd
step the
ta
sk
to pe
rform
is to tra
n
sfo
r
m
the
46.629
+j0.3052
impe
dan
ce i
n
to t
he im
peda
nce seen
by t
h
e
transi
s
to
r as
a DUT
whi
c
h
corre
s
po
nds to 20.
877-j2
5.384O
hm. This impe
dan
ce matchin
g
is
reali
z
ed by u
s
ing the Smith cha
r
t. After
the circui
t opt
imization a
s
we ca
n se
e it on the Figure 8,
the output is
pretty well m
a
tche
d.
The final desi
gned outp
u
t netwo
rk i
s
sh
own at
the Figure 9. The i
nput netwo
rk matchin
g
we
de
signe
d
use
s
th
e
sa
me techniq
u
e
s
a
s
perfo
rm
ed a
bove
but
this i
s
don
e
after the
outp
u
t
netwo
rk a
nd
the loa
d
are
all conn
ecte
d
to the
tra
n
si
stor outp
u
t p
o
rt. Th
e Fig
u
re
10
sh
ows
the
input n
e
two
r
k ci
rcuit of thi
s
de
sign. Sim
u
lated
output
po
wer gai
n v
s
. fre
que
ncy
offset an
d fa
ctor
of stability vs. frequen
cy are pre
s
e
n
ted
in t
he Figure 11 and sh
ows that
the obtaine
d gai
n is
12.972
dB wh
en the rea
c
h
a
b
le maximum
gain is 14.4
6
8dB.
So the o
p
timization
of the
micro-strip
e
li
ne
pa
ram
e
ters mu
st b
e
pe
rforme
d in
order t
o
increa
se the
gain of this p
o
we
r amplifie
r.
Figure 9. The
Matched
Out
put Ne
twork
Figure 10. Input Network
Figure 11. Power
Gain vs.
Freq
uen
cy Offs
et and Fa
ct
or of Stability
vs. Frequ
en
cy
The S param
eters
simulati
on re
sults aft
e
r
optimizatio
n are given b
y
the Figure 12 and
the final desi
gned
circuit is given in the Figure 13.
Evaluation Warning : The document was created with Spire.PDF for Python.
ISSN: 23
02-4
046
TELKOM
NI
KA
Vol. 12, No. 10, Octobe
r 2014: 716
0
– 7166
7166
Figure 12. S11 and S21 Si
mulation
Res
u
lts
after
Optimiz
a
tion
Figure 13. Th
e Final De
sig
ned Ci
rcuit
3. Conclu
sion
Based
on
sim
p
lified cl
ass F
archite
c
ture t
h
ird
-
ha
rmo
n
ic pea
king tu
rni
ng mod
e
was use
d
to desi
gn
a
PA operating
at 5.8G
Hz u
s
ing
a G
a
N
HE
MT tran
sistor. In thi
s
d
e
sig
n
it is
sh
own
how th
e loa
d
pull an
d source p
u
ll techni
que
s are ve
ry
indispen
sa
bl
e to re
du
ce t
he impa
ct of t
h
e
transi
s
to
r p
a
rasitic elem
ent
s o
n
the
po
wer p
e
rfo
r
ma
n
c
e
of a
switch
ing-m
ode
PA, the imp
o
rta
n
c
e
in the
contro
l of voltage
a
nd
cu
rre
nt waveform to
re
duce the
po
wer
dissipate
d
at the h
a
rm
o
n
ic
freque
nci
e
s
wa
s al
so
pre
s
ente
d
. Tran
smissio
n
li
n
e
s
a
r
e
utilize
d
to de
sign
th
e ha
rmoni
cs
tra
p
netwo
rk ta
kin
g
only three
harm
oni
cs int
o
accou
n
t. The sim
u
lated
maximum o
u
tput power
of
50.98
W with the optimum gain of 14dB is pre
s
e
n
ted. Excellent perf
o
rma
n
ce wa
s achieved ov
er
the ban
dwi
d
th of 10
0M
Hz,
for the
po
we
r variatio
n le
ss tha
n
1dB th
e value
of the
drai
n efficie
n
c
y
remai
n
s hi
gh
er than
50%
.Throu
gh th
e re
sults
obt
ained, we ca
n co
nclu
de t
hat the de
sig
ned
Cla
s
s F PA i
s
suita
b
le fo
r t
he field
s
whe
r
e the
hig
her output
p
o
wer and highe
r efficien
cy
are
t
he
main dem
and
s.
Referen
ces
[1]
T
u
ffy
,
N
e
a
l
. A Simplifi
ed Br
oad
ba
nd D
e
si
gn Meth
od
olo
g
y
for
Lin
eariz
e
d
Hi
gh-Efficie
n
c
y
C
onti
nuo
u
s
Class-F Po
w
e
r
Amplifiers,
IEEE Transactio
n
s on Microw
a
v
e Theory an
d
Techni
ques
. 2
012.
[2]
David M S
n
i
d
e
r
. A
T
heoretica
l
Anal
ys
is a
n
d
Exp
e
rime
ntal
Confirmati
on
o
f
the Optimall
y Loa
ded
an
d
Overdriven RF Po
w
e
r Amplifier. IEEE
Trans. On Electron Devices. ED-14,
[3]
Muthus
w
a
m
y
Venkatar
ama
n
i
.
Efficienc
y Improv
em
ent of
W
CDMA Base
Station T
r
ansmitters usin
g
Class-F
p
o
w
e
r
ampl
ifiers. M.S. thesis, De
pt. Elect. Eng
., Virgin
ia P
o
l
y
tec
hnic Institut
e a
nd State
Univ.
,
Blacksburg, Vir
g
inia. 2004.
[4] Kazuhiko
Honjo.
Ultra High
Efficiency Micr
ow
ave Pow
e
r Amp
lifier for
Wireless Pow
e
r Transmiss
ion
.
Procee
din
g
s of
the 42
nd
Europ
ean Micr
o
w
av
e
Confere
n
ce, A
m
sterdam, Net
herl
ands. 2
012
.
[5]
SC Cripps. RF
Po
w
e
r Amp
lifie
rs for W
i
reless
Commun
i
cati
o
n
s.Artech Hou
s
e, Nor
w
o
od, MA. 1999.
[6]
Deh
ao W
u
,
E
Korolki
e
w
i
cz,
Q Lu,
LLi
u.
T
o
Des
i
gn
a
n
d
M
ode
l a
C
l
ass
F
Ampl
ifier
an
d
Investigate
the
Effect of Losses on th
e Effic
i
ency
of dc to
ac Pow
e
r Co
n
v
ersio
n
.
Com
m
unic
a
tion S
ystems Net
w
ork
s
and D
i
gita
l Sig
nal Proc
essin
g
(CSNDSP), 7th Internatio
na
l S
y
mp
osi
u
m on
. 2010.
[7]
Joon
H
y
u
n
g
Ki
m, G
w
e
o
n
Do
Jo, Jun
g
H
o
o
n
Oh, You
n
g
H
oon
Kim, K
w
a
ng
Chu
n
Lee,
Jae H
o
J
u
n
g
.
Mode
lin
g an
d
Desig
n
Meth
o
dol
og
y of Hi
gh
-E
fficienc
y Cl
a
ss-F
and Cl
as
s-F
Po
w
e
r Amplifiers
. IEEE
T
r
ans. On micr
ow
ave theory a
nd techn
i
q
ues
. 201
1; 59(1).
5.
2
5
.
4
5.
6
5
.
8
6.
0
6
.
2
6.
4
6
.
6
6.
8
5.
0
7.
0
-3
0
-2
0
-1
0
0
10
-4
0
20
f
r
eq,
G
H
z
d
B
(S(2
,1
)
)
m1
d
B
(S(1
,1
)
)
m1
fr
e
q
=
dB
(S
(2,
1
))
=
14.
20
9
5.
80
0
G
H
z
MC
R
O
S
O
Cr
o
s
1
W
4
=
2
9
.
69
18
1 mm
W
3
=
1
.
424
56
mm
W
2
=
1
.
094
88
mm
W1
=
0
.
1
m
m
Su
b
s
t
=
"
M
S
u
b
1
"
MT
E
E
_
A
D
S
Te
e
3
W
3
=
50.
30
4 mm
W
2
=
0
.62
5
m
m
W
1
=
0
.
0
00
592
mm
S
ubs
t
=
"
M
S
u
b
1
"
MT
E
E
_
A
D
S
T
ee2
W
3
=
1
.49
4
m
m
W
2
=
0
.
0
00
59
2 mm
W
1
=
1
.
3
12
4 mm
S
u
b
s
t
=
"
M
S
u
b1"
MT
E
E
_
A
D
S
Te
e
1
W3
=
1
.
5
8
9
6
m
m
W2
=
1
.
3
1
2
4
m
m
W
1
=
0
.62
5
m
m
S
u
b
s
t
=
"
M
S
u
b1"
Ma
x
G
a
i
n
Ma
x
G
a
i
n
1
M
a
x
G
a
i
n1
=
m
ax
_g
ai
n
(
S
)
Ma
x
G
a
i
n
MLO
C
TL
4
L=
3.5
2
7
m
m
{
-
t
}
W
=
1.
5
8
9
6
mm {
-
t
}
Su
b
s
t
=
"
M
Su
b
1
"
V_
D
C
S
RC1
Vd
c
=
VG
S
ML
I
N
TL
2
2
L
=
2.5 m
m
W
=
0.
625
mm
Su
b
s
t
=
"
M
S
u
b
1
"
ML
I
N
TL
2
1
L=
2.
5
mm
W
=
0.
62
5 mm
Su
b
s
t
=
"
M
S
u
b
1
"
MLI
N
T
L20
L
=
2.
5 mm
W
=
0.62
5
m
m
Su
b
s
t
=
"
M
S
u
b
1
"
Te
rm
Te
rm
1
Z=
5
0
O
h
m
Nu
m
=
1
MLI
N
TL
1
9
L=
2.5
m
m
W
=
0.
6
25 m
m
Su
b
s
t
=
"
M
S
u
b
1
"
MLS
C
TL
6
L=
6.
3
4
1
mm
{
-
t
}
W
=
50
.30
4
m
m
{
-
t}
S
ubs
t
=
"
M
S
u
b
1
"
ML
S
C
TL
7
L=
4.
5
5
4
7
mm
{
-
t
}
W
=
1.
49
4 mm
{
-
t
}
Su
b
s
t
=
"
M
S
u
b
1
"
ML
O
C
TL
1
6
L=
8.
3
9
7
8
9
m
m
{
-
t
}
W
=
1
.
09
48
8 mm
{
-
t
}
Su
b
s
t
=
"
M
S
u
b
1
"
ML
S
C
TL
1
8
L=
6
.
0
884
9 mm {
-
t
}
W
=
2
9
.6
91
81
m
m
{
-
t
}
Su
b
s
t
=
"
M
S
u
b
1
"
M
C
URV
E
2
Cu
r
v
e
1
N
m
od
e=
2
R
a
di
u
s
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2
.5
m
m
A
n
g
l
e=
90
W
=
0.62
5
m
m
Su
b
s
t
=
"
M
S
u
b
1
"
MC
U
R
V
E
2
Cu
r
v
e
2
Nm
o
d
e
=
2
R
a
di
us
=
2
.5 m
m
A
n
gl
e=
9
0
W
=
0
.
62
5 m
m
Su
b
s
t
=
"
M
Su
b
1
"
MLI
N
T
L17
L
=
13
.
7
628
4 mm {
-
t
}
W
=
0
.
1
000
00
04
815
mm
{
-
t
}
Su
b
s
t
=
"
M
S
u
b
1
"
D
A
_LC
B
a
n
dpa
s
s
S
T
1_
un
t
i
t
l
e
d
2
6
D
A
_LC
B
a
n
dpa
s
s
S
T
1
M
a
x
R
e
a
l
i
za
t
i
o
n
s=
2
5
Rg
=
5
0
O
h
m
T
e
r
m
in
a
t
io
n
=
S
h
o
r
t
T
e
r
m
in
a
t
io
n
R
e
s
p
o
n
s
e
T
y
pe
=
C
he
by
s
h
e
v
N=
2
As
=
2
0
d
B
Ap
=
3
d
B
Fs
2
=
7
G
H
z
Fp
2
=
6
G
H
z
F
p
1
=
5.
6 G
H
z
Fs
1
=
5
G
H
z
S
T
S_
Pa
r
a
m
SP
1
St
e
p
=
1
0
0
M
H
z
St
o
p
=
2
0
G
H
z
S
t
ar
t
=
5 G
H
z
S-
PAR
AM
E
T
E
R
S
Zi
n
Zi
n
1
Z
i
n1
=
z
i
n
(
S
1
1
,
P
or
t
Z
1
)
Zi
n
N
MS
U
B
MS
u
b
1
R
o
ug
h=
0
m
m
T
anD
=
0
T
=
0.
03
5 mm
H
u
=
1
.
0
e+
033
mm
Co
n
d
=
1
.
0
E
+
5
0
Mur
=
1
Er
=
4
.
4
H=
0
.
8
m
m
MS
u
b
DC_
B
l
o
c
k
DC_
B
l
o
c
k
1
DC
_
B
l
o
c
k
D
C
_
B
l
o
ck2
GR
M
1
8
C4
P
a
r
t
Nu
m
b
e
r
=
G
RM
1
8
8
5
C1
H1
0
0
J
A
0
1
GR
M
1
8
C2
P
a
r
t
N
u
m
b
er
=
G
R
M
18
5R
61C
1
05K
E
4
4
GR
M
1
8
C3
P
a
r
t
Nu
m
b
e
r
=
G
RM
1
8
8
5
C1
H8
2
0
J
A
0
1
GR
M
1
8
C1
P
a
r
t
N
u
m
b
er
=
G
R
M
18
85
C
1
H
8
20
J
A
01
MU
R
A
T
A
I
n
c
l
u
d
e
mu
R
a
t
a
NE
T
L
I
S
T
I
N
CL
UD
E
VA
R
VA
R
1
VG
S
=
-
2
.
1
V
VD
S
=
2
8
V
Eq
n
Va
r
V_
D
C
SR
C
2
Vd
c
=
V
D
S
ML
I
N
TL
2
L
=
8.44
5
m
m
W
=
1.
494
mm
Su
b
s
t
=
"
M
S
u
b
1
"
ML
I
N
TL
1
L=
3.
8
3
mm
W
=
1
.
49
4 mm
Su
b
s
t
=
"
M
S
u
b
1
"
MLI
N
TL
1
5
L=
9.2
0
4
m
m
{
-
t
}
W
=
1
.
42
45
6 m
m
{
-
t}
S
u
b
s
t
=
"
M
S
ub1
"
MLI
N
TL
3
L
=
1
1
.
4
97
6 mm {
-
t
}
W
=
1
.
31
24
m
m
{
-
t}
Su
b
s
t
=
"
M
S
u
b
1
"
Te
r
m
Te
r
m
2
Z=
5
0
O
h
m
Nu
m
=
2
MLI
N
TL
5
L=
15
.46
9
m
m
{
-
t}
W
=
0.
0
005
92
mm
{
-
t
}
Su
b
s
t
=
"
M
S
u
b
1
"
C
G
H
4
0
0
45
F
_
r
4
a
_
c
r
ee
_p
ac
k
a
g
e
_
4
0
_
r
5
X2
cr
t
h
=
2
.
7
t
c
as
e=
25
C
r
ee C
G
H
400
45F
St
a
b
F
a
c
t
St
a
b
F
a
c
t
1
S
t
a
b
F
a
c
t
1=
s
t
ab
_f
ac
t
(
S
)
St
a
b
F
a
c
t
Pw
r
G
a
i
n
Pw
r
G
a
i
n
1
P
w
r
G
ai
n
1
=
p
w
r
_g
ai
n
(
S
,
P
o
r
t
Z
1
,
P
o
r
t
Z
2
)
Pw
r
G
a
i
n
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